Discrete multi-tone systems for half-duplex ip links

ABSTRACT

A DMT system for a half-duplex two-way link carries Internet protocol encoded video stream on a coaxial cable that also carries a baseband rendition of the same video stream. A plurality of downlink symbols modulated on a subband of subcarriers in a downlink signal are decoded. The symbols may carry data encoded on a subband using a constellation of QAM symbols assigned to the subband. Other subbands may be associated with different QAM constellations. Lower-order constellations of QAM symbols may be assigned to subbands that include higher-frequency subcarriers and higher-order constellations of QAM symbols may be assigned to subbands that include lower-frequency subcarriers. A block error correction decoder may be synchronized based on an identification of the first constellation of QAM symbols and information identifying boundaries between the plurality of downlink symbols.

CROSS-REFERENCE TO RELATED APPLICATION(S)

This application is a continuation of, and claims benefit of priorityfrom U.S. patent application Ser. No. 13/706,290, filed on Dec. 5, 2012,and which claimed the benefit of U.S. Provisional Application Ser. No.61/589,101, entitled “Discrete Multi-tone Systems And Methods ForHalf-Duplex IP Links” which was filed on Jan. 20, 2012, and both ofthese applications are expressly incorporated by reference herein intheir entirety.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates frequency response of 2000 feet of RG-59 cable.

FIG. 2 illustrates frequency tilt over 2000 feet of RG-59 cable in asignal band used in certain embodiments of the invention.

FIG. 3 is a frequency distribution chart illustrating an example ofconstellation to subband assignment according to certain aspects of theinvention.

FIG. 4 is a schematic drawing and a spectral occupancy chartillustrating one example system according to certain aspects of theinvention.

FIG. 5 is a schematic illustrating a camera side modem according tocertain aspects of the invention.

FIG. 6 is a schematic illustrating a DVR example of a downside sidemodem according to certain aspects of the invention.

FIG. 7 is a schematic illustrating a DMT transmitter according tocertain aspects of the invention.

FIG. 8 is a diagram illustrating an interleaver and a deinterleaver.

FIG. 9 is a randomizer according to certain aspects of the invention.

FIG. 10 is a schematic illustrating a punctured trellis code modulatoraccording to certain aspects of the invention.

FIG. 11 is a schematic illustrating QAM constellations.

FIG. 12 is a schematic illustrating half-duplex operation and DMTspectrum according to certain aspects of the invention.

FIG. 13 is a chart illustrating side lobe energy reduction for thedownstream signal according to certain aspects of the invention.

FIG. 14 illustrates discrete time-windowed half-band filter tapsaccording to certain aspects of the invention.

FIG. 15 illustrates a half-band filter response according to certainaspects of the invention.

FIG. 16 illustrates DMT magnitude spectra related to sample rateconversion.

FIG. 17 illustrates probabilities of one or more clips within one DMTsymbol.

FIG. 18 is a plot illustrating spectral densities of DMT signal, hardclip noise, clip noise and significant out-of-band noise reductionaccording to certain aspects of the invention.

FIG. 19 is a schematic representation of a frequency domain filteringoperation according to certain aspects of the invention.

FIG. 20 is a schematic illustrating half-duplex alternating downstreamand upstream DMT symbols according to certain aspects of the invention.

FIG. 21 is a flowchart of a method for determining confidence accordingto certain aspects of the invention.

FIG. 22 is a flowchart of a method for synchronizing transmitter andreceiver counters according to certain aspects of the invention.

FIG. 23 illustrates an impulse duration tolerance goal according tocertain aspects of the invention.

FIG. 24 is a chart illustrating frequency response for 3000 feet ofRG-59.

FIG. 25 is a frequency distribution chart illustrating an example ofconstellation to subband assignment according to certain aspects of theinvention.

FIG. 26 is a schematic illustrating a DMT receiver according to certainaspects of the invention.

FIG. 27 illustrates sample clock synchronization and side receiver rateconversion according to certain aspects of the invention.

FIG. 28 is a high-level flow diagram of the Reed-Solomon decoderaccording to certain aspects of the invention.

FIG. 29 illustrates analog front ends according to certain aspects ofthe invention.

FIG. 30 shows a frequency response of a filter used in certainembodiments of the invention.

FIG. 31 is a simplified block schematic illustrating processing systemsemployed in certain embodiments of the invention.

FIG. 32 includes flowcharts of methods for communication according toaspects of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Embodiments of the present invention will now be described in detailwith reference to the drawings, which are provided as illustrativeexamples so as to enable those skilled in the art to practice theinvention. Notably, the figures and examples below are not meant tolimit the scope of the present invention to a single embodiment, butother embodiments are possible by way of interchange of some or all ofthe described or illustrated elements. Wherever convenient, the samereference numbers will be used throughout the drawings to refer to sameor like parts. Where certain elements of these embodiments can bepartially or fully implemented using known components, only thoseportions of such known components that are necessary for anunderstanding of the present invention will be described, and detaileddescriptions of other portions of such known components will be omittedso as not to obscure the invention. In the present specification, anembodiment showing a singular component should not be consideredlimiting; rather, the descriptions herein are intended to encompassother embodiments including a plurality of the same component, andvice-versa, unless explicitly stated otherwise. Moreover, applicants donot intend for any term in the specification or claims to be ascribed anuncommon or special meaning unless explicitly set forth as such.Further, embodiments of the present invention encompass present andfuture known equivalents to the components referred to herein by way ofillustration.

As described herein, a first generation security link over coax (SLOC)may be characterized as a full-duplex, single-carrier, point-to-point,2-way digital video over IP link with concurrent and/or simultaneoustransmission of analog composite video in the downstream direction,blanking, and sync (CVBS) over coaxial cable for security applications.In one example, a first generation SLOC system may provide a downstreambit rate of 36 megabits per second (Mbps) from camera to monitor and/ordigital video recorder (DVR) over 1000-1500 feet of low-cost RG-59cable. Advanced Internet protocol (IP) cameras can output data atsignificantly higher bit rates which can extend beyond 36 Mbps.

Certain embodiments of the invention employ a discrete multi-tone (DMT)system in a half-duplex, bidirectional IP data link to provide anext-generation SLOC. In the downstream direction (camera to DVR),Internet protocol data (usually compressed digital video) may betransmitted in combination with CVBS. Internet protocol acknowledgepackets may be sent in the upstream (DVR to camera) direction. The nextgeneration SLOC system can support 100 Mbps downstream and 6.25 Mbpsupstream over at least 2000 feet of low cost RG-59 cable with the bitrate automatically adaptively reduced as the cable gets longer. At 3000feet a bit-rate of 52.3 Mbps downstream and 3.26 Mbps upstream can beachieved. An upstream link may be provided from monitor and/or DVR tothe camera. Simultaneous CVBS transmission in the downstream directionmay be accommodated by zeroing the digital spectrum in the range of DCto 11 MHz, for example.

A simple and effective bit-loading (BL) method can be used to exploitthe cable channel capacity. The BL method allows the system to overcomethe substantial frequency tilt in long sections of cable (see FIG. 1),resulting in a high bit rate and/or long cable reach. A data framingmethod disclosed herein, closely coupled with the BL method,Reed-Solomon packetization (or the like) and byte interleaving, may beused to map the input data to the modulation in a manner that providesfor simple synchronization of deinterleaving and decoding components ina receiver. The decoding components may comprise a block-errorcorrecting decoder, such as the Reed-Solomon decoder described herein.Synchronization may operate in a consistent manner regardless ofbit-loading assignments. Controlled clipping of the up-sampled signal atthe transmitter may reduce the peak-to-average power ratio and mayimprove transmitted SNR, while limiting the generation of in-band noise.Raised-cosine windowing of the up-sampled signal controls out-of-bandnoise to prevent digital interference into the CVBS signal, usingtechniques and methods known in the art. Use of a small number of movingpilots allows for effective channel estimation, enabling trackingdigital adaptive equalization for the DMT signal. The DMT channelestimator may be used to estimate the frequency tilt in the CVBS band.This information can be used to enable effective analog or digitalequalization of the CVBS signal. A low-noise AFE is described thatenables certain systems to meet performance goals.

FIG. 1 is a graph 100 showing a typical frequency response for 2000 feetof RG-59 cable. In a single-carrier system, all transmitted digitalquadrature amplitude modulation (QAM) symbols are typically members ofthe same constellation set. For high bit rates, high-orderconstellations may be needed to encode more bits of information persymbol. High-order constellations in turn require relatively highreceiver SNRs for reliable reception. The substantial high-frequencyattenuation makes this difficult to achieve in conventional systems. Ifthe constellation order is reduced, however, a yet wider signalbandwidth may be needed with greater attenuation occurring at the highend of the enlarged signal band.

Certain embodiments employ DMT (baseband multicarrier) modulation, inwhich a signal comprises a set of subcarriers. Each of a plurality ofsubbands of the subcarriers may be modulated by QAM symbols from aconstellation selected for, and/or matched to the subband.Lower-frequency subcarriers typically suffer less attenuation thanhigher frequency subcarriers, and lower-frequency subcarriers aretypically received with a higher post-analog front-end (AFE)signal-to-noise ratio (SNR) than are the higher-frequency subcarriers.Accordingly, lower-frequency subcarriers can reliably supporthigher-order constellations. To provide reliable reception of thehigher-frequency subcarriers, the higher-frequency subcarriers aremodulated by symbols from lower-order constellations. With thisarrangement, the lower-frequency subcarriers can support a higher bitrate per subcarrier than do the higher-frequency subcarriers. Overall,this scheme allows the system to achieve a bit-rate/reliabilitytrade-off that approaches more closely the channel capacity in than canbe practically achieved with a single-carrier system.

FIG. 2 is a graph 200 showing a typical frequency response for 2000 feetof RG-59 cable for a subset of bandwidth 202 corresponding to a SLOCdigital signal. The band of frequencies 202 in a SLOC digital signal canextend from about 11 MHz to about 42 MHz and, as shown in the graph 200,a 2000-foot length of RG-59 cable can cause about 20 dB of tilt acrossthis band 202. While some embodiments may provide differentconstellation assignments on an individual subcarrier basis, goodperformance for SLOC over coaxial cable can be achieved with lessgranularity in constellation assignment. The baseband 204, shown asbetween 0 Hz and 11 MHz, may be reserved for carrying a simultaneousanalog video signal. The baseband signal may be equalized at thereceiver to correct for phase shifts and attenuation.

FIG. 3 is a chart 300 illustrating an example of constellationassignment for a specific example of subband configuration. In theexample, available bandwidth in a signal is divided into 44 subbands, asillustrated by subband 302 for example. In the example, the availablebandwidth excludes CVBS bandwidth 304. The subbands may be of equalbandwidth and may comprise an integer number of adjacent subcarriers.Subbands may be divided into groups 306 a-306 g and QAM constellationsmay be assigned to the groups 306 a-306 g. The groups 306 a-306 g maycomprise different numbers of subbands. In the example depicted in FIG.3, each subband in group 306 a is assigned a 512 QAM constellation, eachsubband in group 306 b is assigned a 256 QAM constellation, each subbandin group 306 c is assigned a 128 QAM constellation, each subband ingroup 306 d is assigned a 64 QAM constellation, each subband in group306 e is assigned a 32 QAM constellation, each subband in group 306 f isassigned a 16 QAM constellation and each subband in group 306 b isassigned a 8 phase shift keying (PSK) constellation.

Certain embodiments using the QAM constellations assigned per FIG. 3,may additionally use punctured trellis coded modulation to transmitReed-Solomon-coded data can result in quasi-error-free (QEF) SNRthresholds at the receiver trellis decoder input that range from 37.8 dBfor 512 QAM to 8.8 dB for 8 PSK. In the chart 300, y-axis value (height)of each group 306 a-306 g indicates its respective QEF threshold. Thisspan of SNR thresholds closely approximates the cable tilt and eachgroup 306 a-306 g can have approximately the same SNR margin at thereceiver. QAM constellations may be automatically assigned to groups 306a-306 g using any appropriate or adaptive processes and certainwell-known techniques may produce near-optimal results in terms ofapproaching channel capacity given the coding and modulation utilized ina system.

FIG. 4 is diagram 400 illustrating a SLOC system according to certainaspects of the present invention. A SLOC-equipped camera 402 may beconnected by coaxial cable 404 to a SLOC-equipped digital video recorderDVR 408 or other receiving equipment such as video servers and videorouters, etc. SLOC-equipped camera 402 comprises a modem 414 thattransmits one or more signals representative of a video stream generatedor otherwise processed and/or relayed by multi-media processor 412. Theone or more signals may include an analog CVBS signal 416 transmitted asa baseband signal, and a concurrently transmitted digital DMT signal 418occupying the band from 11.2 MHz to 42.3 MHz which carries data suppliedby a media independent interface (MII), as illustrated graphically at440. The DMT signal accommodates the MII internet protocol signal 418that typically carries compressed high definition digital video in thedownstream direction to the DVR at nominal rates of up to 100 Mbps. TheCVBS signal 416 may carry a standard definition version of thecompressed high definition digital video carried in the DMT signal.

The DVR 408 may also comprise a modem 424 that receives and separates asignal received from the coaxial cable 404 into a received CVBS signal426 and a received downstream DMT signal portion of MII 428. Thereceived downstream DMT signal is decoded and passed via an MIIinterface to a host processor 430 which decodes, or causes to bedecoded, the compressed digital high definition video in the MII signal428. Modem 424 may also transmit an upstream DMT signal which carriesreturn data from the MII signal 428. The upstream DMT signal may betransmitted in a half-duplex manner at bit rates of up to 6.25 MHz. Theupstream DMT signal may encode an audio feed, camera control informationand/or other information, such as messages required to maintain anInternet protocol connection between the upstream camera 402 anddownstream DVR 408 or other devices. In one example, the upstream DMTsignal occupies the frequency band from 11.07 MHz to 43.19 MHz, which isnearly the same as the frequency band used by the downstream DMT signal.

FIGS. 5 and 6 include schematics 500 and 600 respectively, which presenthigh-level views of camera side SLOC modem 502 and DVR side SLOC modem602. Camera side modem 502 receives an IP MII signal 530 which is fed todownstream DMT transmitter 520. DMT transmitter 520 may provide anominal 100 Mbps output to digital-to-analog converter (DAC) 514, whichconverts the signal to an analog signal and is combined with thecamera's CVBS signal 532 in adder 528. Adder 528 may drive the combinedsignal onto the cable 514. AFE 516 includes a band pass filter (“BPF”)518 to reduce interference added into the CVBS at the low end and tosuppress the DMT image at the high end. The lower bit rate upstream DMTsignal can be received from the cable 514 and fed to camera side DMTreceiver 510 for decoding. AFE 504 may comprise a band pass filter (BPF)506 attenuates the CVBS signal while passing the upstream DMT signalbefore providing the DMT signal for digitization by analog-to-digitalconverter (ADC) 508. An output of DMT receiver 510 may be provided tothe MII interface 534.

At the DVR side SLOC modem 602, the combined CVBS/DMT signal receivedfrom cable 610 is fed to AFEs 604, 614, and 624. AFE 614 filters outmost of the CVBS and feeds the filtered DMT signal to the downstream DMTreceiver 620 after digitization by ADC 618. In the CVBS signal path, theLPF of AFE 624 filters out most of the DMT signal. In some embodiments,an ADC, digital LPF, and DAC can be inserted for further DMTsuppression. AFE 604 has a post-DAC 608 BPF 612 similar to BPF 518 ofthe camera side modem 502. An analog CVBS equalizer 610 providescompensation for high-frequency roll-off attributable to the cable.Return packets emanating from the host MII interface are transmittedupstream by the upstream DMT transmitter 606, which is structurallysimilar to the downstream DMT transmitter 520, but may be configuredwith different operating parameters as appropriate or desired. DAC 608converts the upstream DMT signal to an analog signal for transmissionthrough AFE 604, which includes BPF 612.

SLOC DMT Transmitter

FIG. 7 is a block schematic 700 that illustrates a DMT transmitteraccording to certain aspects of the invention. An MII transmit interface(MIITx) 702 receives the MII data. Data from the MIITx is formed intobytes and then into packets of size k bytes at the input to aReed-Solomon (RS) coder 704, which calculates and appends r=24 paritybytes to the packet. The resulting packet size P=k+24 can vary accordingto the subcarrier constellation assignments in a manner to be describedlater. Up to t=r/2=12 corrupted bytes per packet can be corrected at thereceiver. The RS coder output is then processed by a convolutional byteinterleaver 706. FIG. 8 illustrates an implementation 800 of interleaver706. Interleaver 706, along with a complimentary deinterleaver 850 inthe SLOC receiver 620, may be used in any suitable manner s known in theart, and may be configured to combat impulse noise affecting thetransmitted signal. This impulse noise can couple into the systemthrough a power line, or from the very long lengths of coaxial cable,which can pick up electrical noise even though the coax cable isshielded. Such impulse noise can cause IP packet loss.

According to certain aspects of the invention, a noise impulse having aduration of one “package” can be tolerated, where the content and sizeof the package is discussed in more detail below. Structures 800 and 850comprise B shift registers of increasing size. For interleaver 800, thetopmost shift register 802 has length zero, and the bottommost shiftregister 804 has length (B−1) M bytes where M is typically a smallinteger, and typically BM≧P. For both the interleaver 800 anddeinterleaver 850, a synchronization signal may be needed to force inputcommutator 806 and output commutator 808 to the top position at the samepoint in the byte stream, and to synchronize the interleaving to thedeinterleaving. This synchronization is discussed in more detailelsewhere herein. The input and output commutators 806, 808 move downone position as a byte enters the interleaver and a different byte exitsthe interleaver. When the commutators 806, 808 reach the bottom, theyshift back to the top. The RS coder 704/interleaver 706 combinationallows a corresponding deinterleaver/RS decoder in the receiver tocorrect data corrupted by noise impulses of duration Bt bytes.

Output bytes of interleaver 706 are then randomized by randomizer 708 toensure a uniform distribution of QAM symbols. FIG. 9 shows an example ofa downstream randomizer 708 based on the polynomial:

x ¹⁶ +x ¹³ +x ¹² +x ¹¹ +x ⁷ +x ⁶ +x ³ +x+1

In the example described herein, the upstream randomizer is the same asthe downstream randomizer 708, except that the 14^(th) and 15^(th)stages 902 and 904 are removed.

The data is then encoded using punctured trellis code modulation (PCTM).FIG. 10 is a diagram 1000 illustrating a PTCM coder 712. The PTCM coderencodes m−1 bits at a time from the randomizer into m bits. The code maybe based on a ½ rate mother code. Bits output by the ½ rate coder maythen be deleted (punctured) according to a specified pattern to producehigher rate (m−1)/m rate codes. The quantity m may range from m=2 for4-QAM symbols to m=9 for 512-QAM symbols. The QAM constellation numberis given by 2 m. The assignment of a particular QAM constellation to aparticular DMT subband of subcarriers (the selection of m) is discussedin more detail below.

The m data bits are then mapped by mapper 714 to points in the selected2^(m) QAM constellation. The 256-, 64-, 16- and 4-point constellationsare square. The 128- and 32-point constellations are crossconstellations. These constellations may be scaled so that all have thesame average power, assuming a uniform distribution of symbols. FIG. 11shows three possible constellations 1100, 1110 and 1120 plotted in thecomplex plane. Complex QAM data symbols are fed to the QAM symbolmultiplexer (mux) 716. The mux 716 may also input fixed-level BPSKtiming sync and pilot symbols, which are multiplexed with the QAM datasymbols, all of which are assigned to certain DMT subcarriers withineach DMT data symbol.

The mux 716 periodically inserts a superframe DMT sync symbol at regularintervals via the special input from module 718. This DMT symbol maycomprise subcarriers modulated by a specific uncoded binary phase shiftkeyed (BPSK) pseudorandom number (PN) sequence which is easilydetectable by the receiver. This enables receiver synchronization ofcertain processes that will be discussed later. Just after the DMTsuperframe sync symbol, either two (upstream) or one (downstream) DMTsystem data (sysdata) symbols are inserted, also via the special inputfrom module 718. These symbols carry the constellation to subcarrierassignments and other critical parameters and information. The contentsand structure of the DMT sysdata symbols are also further discussedelsewhere in this disclosure.

During system start-up, a series of DMT symbols that comprise onlyun-coded BPSK training symbols derived from two other PN sequences (tr0,tr1) are transmitted to assist receiver timing synchronization andinitial channel estimation. Details of system start-up are discussedherein below.

DMT Modulation Parameters

As a first step in the downstream DMT modulation, a 4 k-point realoutput of IFFT 720 is obtained by creating a complex conjugate inputsequence from a 2 k-long input data block. The upstream transmitter mayutilize a 512-point real output IFFT. The nominal sample frequency isF_(s)=90 MHz. A 2 μs (180 samples) cyclic suffix (CS) is appended to theIFFT output vector to form one DMT symbol. The CS allows the receiver toresist inter-DMT-symbol interference and greatly simplifies digitalequalization. The resulting DMT parameters are shown in Table 1.

TABLE 1 IFFT symbol time samples per size with CS in μs DMT sym binwidth Hz downstream 4096 D*/2 = 47.51 4276 21972.66 upstream 512 U* =7.69 692 175781.25

Cable propagation delay results in added overhead in a half-duplexsystem. For 1000 meters of cable, the one-way delay (OWD) is about 5 μs.With some added safety margin, an allowance for a maximum OWD of 6 μs(540 samples) is made, or a round-trip delay (RTD) of 12 μs. Thehalf-duplex operation is illustrated in the timing chart 1200 of FIG.12, as viewed from the camera side and DVR side of the cable. The linealternates between two DMT symbols 1202, 1204 sent downstream and oneDMT symbol 1206 sent upstream. Each of D1 1202, D2 1204 and D3 1208represents a pair of downstream DMT symbols of duration 2×47.51 μs=95.02μs. Each of U1 1206 and U2 1210 is a single upstream DMT symbol ofduration 7.69 μs. The combination of two downstream symbols followed byone upstream symbol and the intervening space is referred to herein as a“package.”

In one example, 2048 downstream subcarriers are assigned as follows

-   -   null subcarriers providing a hole for CVBS:        -   0-509 (510 subcarriers, 0-11.20 MHz)    -   useful subcarriers:        -   510-1923 (1414 subcarriers)    -   null subcarriers to prevent aliasing (guard bands):        -   1924-2047 (124 subcarriers), about 6% of DMT band

Of the useful subcarriers, four are permanently modulated with a fixedBPSK symbol and utilized to assist receiver timing synchronization.These are subcarriers 1025, 1041, 1057, and 1073. Two pilot subcarriersmodulated with a fixed BPSK symbol are also utilized to assist receiverequalization. This pair of subcarrier frequencies change for each DMTsymbol according to a known pattern, rotating or otherwise circulatingthrough all the even useful subcarriers as well as odd subcarrier 1923.This leaves 1408 subcarriers available to carry IP data.

In one example, 512 upstream subcarriers are assigned as follows:

-   -   null subcarriers providing a hole for CVBS:        -   0-62 (63 subcarriers, 0-11.07 MHz)    -   useful subcarriers:        -   63-239 (177 subcarriers)    -   null subcarriers to prevent aliasing:        -   240-255 (16 subcarriers), about 6% of DMT band

Of the useful subcarriers, one is used for a moving pilot subcarriermodulated with a fixed BPSK symbol that aids receiver equalization. Thissubcarrier frequency changes for each DMT symbol according to a knownpattern, rotating or otherwise circulating through all the usefulsubcarriers. The remaining 176 subcarriers are available to transmit IPdata.

The resulting ideal DMT spectrum 1220 is shown in FIG. 12. For thedownstream signal, f₁=11.20 MHz and f₂=42.28 MHz. For the upstreamsignal, f₁=11.07 MHz and f₂=42.19 MHz.

DMT Modulation Description

For IFFT: Let A[n] be the mth N-length vector of QAM symbols at theinput to the N-point IFFT module of FIG. 8. Let that vector be given by

${A\lbrack m\rbrack} = {\left\lbrack {{A_{\frac{- N}{2}}\lbrack m\rbrack}{A_{\frac{- N}{2} + 1}\lbrack m\rbrack}\mspace{14mu} \ldots \mspace{14mu} {\ldots.\mspace{20mu} {A_{\frac{N}{2} - 1}\lbrack m\rbrack}}} \right\rbrack.}$

Then the IFFT efficiently computes the vector

${x\lbrack m\rbrack} = \left\lbrack {{x_{\frac{- N}{2}}\lbrack m\rbrack}{x_{\frac{- N}{2} + 1}\lbrack m\rbrack}\mspace{14mu} \ldots \mspace{14mu} {\ldots.\mspace{20mu} {x_{\frac{N}{2} - 1}\lbrack m\rbrack}}} \right\rbrack$where${{x_{k}\lbrack m\rbrack} = {\frac{1}{\sqrt{N}}{\sum\limits_{n = {{- N}/2}}^{{N/2} - 1}\; {{A_{n}\lbrack m\rbrack}^{\frac{j\; 2\; \pi \; {kn}}{N}}}}}},{k = \frac{- N}{2}},{\frac{- N}{2} + 1},\ldots \mspace{14mu},{\frac{N}{2} - 1.}$

Note that A[n] is conjugate symmetric, i.e.

${{A_{- n}\lbrack m\rbrack} = {A_{n}^{*}\lbrack m\rbrack}},{n = 1},2,\ldots \mspace{14mu},{\frac{N}{2} - 1},{and}$$A_{\frac{- N}{2}} = {A_{0} = 0.}$

Therefore, vector x[m] is real.

The downstream 4096-point decimation in time (DIT) IFFT comprises sixradix-4 stages. The upstream 512-point DIT IFFT is composed of fourradix-4 stages followed by one radix-2 stage. IFFT/FFT architecture isdiscussed further in the description of the DMT receiver. Note that dueto the operation of the DIT IFFT, the input QAM symbol stream ismodulated by subcarriers numbered in bit-reversed order. Hence, abit-loading table is accessed to assign the correct constellation to thecurrent subcarrier. The IFFT output is in normal subcarrier order.

The system may have a module 722 that inserts a cyclic suffix (CS). Forconvenience the IFFT output vector may be re-indexed as follows

[x ₀ [m]x ₁ [m] . . . x _(N−1) [m]]

In some embodiments, a CS of N_(CS)=180 samples (2 μs) is appended tothe IFFT output vector. This is given as

x _(ext) [m]=[x ₀ [m]x ₁ [m] . . . x _(N−1) [m]x ₀ [m]x ₁ [m] . . . x₁₇₉ [m]].

The CS can be thought of as a time domain guard interval betweensuccessive DMT symbols. If this guard interval is longer than theexpected channel impulse response of the cable, then inter-DMT-symbolinterference is prevented. This interference may be prevented becausethere will then be at least N samples of the received DMT symbol that donot contain any component of another delayed DMT symbol.

The cyclic extension provides another advantage because it allows forthe simple removal of the intra-DMT-symbol interference. When thecircularly extended DMT symbol is linearly convolved with the cablechannel impulse response, the effect as seen by the receiver is as ifthe unextended DMT symbol were circularly convolved with the cablechannel impulse response. Given a good estimate of that response basedon QAM pilot subcarriers, ideally the intra-DMT-symbol interference canbe completely removed with frequency domain adaptive equalization whichis effectively a circular convolution operation. This holds as long asthe channel impulse response is not longer than the CS.

With reference now to windowing module 724, the DMT signal has strongand slowly decreasing side lobes in the frequency domain caused by timedomain waveform discontinuities at the DMT symbol boundaries. This canmake it difficult to remove spectral images when the signal isinterpolated to 2×F_(samp). Also, AFE filtering to prevent DMTinterference into CVBS can be difficult. Windowing is very effective insmoothing the DMT time domain symbol transitions and thus reduces theout-of-band energy in exchange for losing the effectiveness of a smallpart of the CS. In addition to the CS, N_(W) more cyclic samples forwindowing are also appended to x[m] to form:

y[m]=[x ₀ [m]x ₁ [m] . . . x _(N−1) [m]x ₀ [m]x ₁ [m] . . . x _(N) _(CS)_(+N) _(W) ⁻¹ [m]]=[y ₀ [m]y ₁ [m] . . . y _(N+N) _(CS) _(+N) _(W) ⁻¹[m]].

As will be appreciated, raised-cosine windowing of the DMT symbol cansubstantially reduce the DMT signal side-lobes. Windowing is performedon y[m] in the following manner:

{tilde over (y)} _(k) [m]=y _(k) [n]w _(k) ,k=0,1, . . . ,N+N _(Cs) +N_(W)−1  (eqn. 1)

where w_(k) are samples of the raised cosine function given by

$w_{k} = \left\{ \begin{matrix}{{\frac{1}{2}\left\lbrack {1 - {\cos \left( \frac{\pi \; k}{\beta \; N_{s}} \right)}} \right\rbrack},} & {{k = 0},1,\ldots \mspace{14mu},N_{W}} \\{1,} & {{k = {N_{W} + 1}},\ldots \mspace{14mu},{N_{s} - 1}} \\{{\frac{1}{2}\left\lbrack {1 + {\cos \left( \frac{\pi \left( {k - N_{s}} \right)}{\beta \; N_{s}} \right)}} \right\rbrack},} & {{k = N_{s}},\ldots \mspace{14mu},{N_{s} + N_{W}}}\end{matrix} \right.$where N _(s) =N+N _(CS) and β=N _(W) /N _(s).

SLOC uses N_(W)=21 samples.

In DMT systems that use windowing, the final N_(W) samples of each DMTsymbol may be overlapped (summed) with the first N_(W) samples of thenext DMT symbol. This reduces the effective length of the CS by N_(W)samples. However, in certain embodiments of the presently disclosed SLOCsystem, this approach is modified due to the half-duplex nature of thescheme. For the upstream DMT symbols, there is no overlap needed becausethose DMT symbols are always isolated; i.e. there is no adjacent DMTsymbol being transmitted. As shown in constellation 1200 of FIG. 12, thedownstream DMT symbols come in isolated pairs. Hence the final N_(W)samples of the first symbol of the pair overlap the first N_(W) samplesof the second symbol of the pair. The overlapped pair of DMT symbols isgiven by:

$\begin{matrix}{{{\overset{\sim}{z}}_{k}\lbrack m\rbrack} = \left\{ \begin{matrix}{{{\overset{\sim}{y}}_{k}^{(0)}\lbrack m\rbrack},} & {{k = 0},1,\ldots \mspace{14mu},{N_{s} - 1}} \\{{{{\overset{\sim}{y}}_{k}^{(0)}\lbrack m\rbrack} + {{\overset{\sim}{y}}_{N_{s} - k}^{(1)}\lbrack m\rbrack}},} & {{k = N_{s}},\ldots \mspace{14mu},{N_{s} + N_{W} - 1}} \\{{{\overset{\sim}{y}}_{k - N_{s}}^{(1)}\lbrack m\rbrack},} & {{k = {N_{s} + N_{W}}},\ldots \mspace{14mu},{{2\; N_{s}} + N_{W} - 1}}\end{matrix} \right.} & \left( {{eqn}.\mspace{14mu} 2} \right)\end{matrix}$

where the superscripts (0) and (1) designate the first and second DMTsymbols of the pair, respectively.

The side lobe energy reduction for the downstream signal is shown in thechart 1300 of FIG. 13. The 90 MHz sampled output of the windowingoperation, {tilde over (y)}_(k) [m] for upstream or {tilde over (z)}_(k)[m] for downstream is fed to a 2× up-sampling half-band filter.

Some embodiments employ a half-band up-sampling filter 726. Afterwindowing, a next step may be to up-sample the signal of eqn. 1 or eqn.2 by a factor of two. This may be achieved by inserting a zero samplebetween each data sample and then filtering the sequence with ahalf-band filter to remove the spectral images. An ideal infinitely longhalf-band filter has a brick wall frequency response. A practicalimplementation may truncate the discrete-time tap values using a windowfunction which will cause some pass-band ripple and some slope in thepass-band to stop-band region. Tap values 1400 used in the SLOCtransmitter half-band filter are shown in FIG. 14. Note that except forthe center tap, every other tap value is zero. The resulting frequencyresponse 1500 is shown in FIG. 15.

FIG. 16 is a drawing 1600 illustrating signal spectra 1602, 1604, 1606and 1608. Spectrum 1602 illustrates the DMT signal spectrum at the inputto the half-band filter with the sample rate F_(s)=90 MHz. The 2×up-sampled signal spectrum with resultant images is shown in thespectrum 1604. Application of this discrete-time half-band filterresults in the spectrum 1606.

Subsequent post-DAC AFE filtering removes the image near 180 MHz as seenin the spectrum 1608. If the length B half-band filter function is h_(i)^((HB)), then the filter output is

$b_{n} = {\sum\limits_{i = 0}^{B - 1}\; {{\overset{\sim}{z}}_{i}{h_{n - i}^{({HB})}.}}}$

In some embodiments, a clipping module 728 applies a clipping algorithm.For a DMT signal, the peak-to-average power ratio (PAPR) can be veryhigh. The PAPR is defined as

${{PAPR}\lbrack m\rbrack}\overset{\bigtriangleup}{=}{\frac{\max\limits_{k,m}\left\{ {b_{n}b_{n}^{*}} \right\}}{E\left\lbrack {b_{n}b_{n}^{*}} \right\rbrack}.}$

The PAPR is proportional to the number of subcarriers in a DMT symbol. Aconstant of proportionality differs based on the constellationsmodulating each subcarrier. The PAPR may be important because, if onewants to avoid signal clipping, substantial headroom is needed. However,this results in the need for a very large (in terms of number of bits)DAC/ADC to provide sufficient resolution (and thus low quantizationnoise).

FIG. 17 includes a chart 1700 of the probability of the PAPR of a DMTsymbol exceeding the x axis value for the case of a DMT signal modulatedwith random 64 QAM symbols. This is equivalent to the probability of oneor more clips within a DMT symbol if the DAC headroom is set to thecorresponding x axis value. Plots are given for the two IFFT lengthsused in SLOC. The probability of a clip for a given headroom increasesas the IFFT length increases. Such clipping of the half-band filteroutput generates both in-band and out-of-band (OOB) noise. Moreover,while more clips occur for larger IFFTs at a given headroom setting, theclip noise generated is averaged over more samples by the receiver FFToperation. This greatly diminishes the effect of clipping. A spectralplot of clipping noise 1702 with a headroom setting of 12 dB compared tothe 4 k DMT signal 1704 is shown in FIG. 18.

If clipping is executed at the fundamental DMT sample rate (90 MHz forSLOC), the noise generated is in-band. If the signal is clipped after 2×up-sampling, less in-band noise is generated, but significant OOB noisethen appears. Low pass filtering can then reduce the OOB noise but alsocauses some regrowth in PAPR, partially undoing the benefit of clippingin the first place.

One effective method for hard clipping the up-sampled multicarriersignal and filtering out the OOB noise can be executed in an iterativemanner. Since the filtering causes some PAPR regrowth, the method can beexecuted in an iterative manner. More iterations may achieve bettercontrol of the PAPR while at the same time creating more in-band noise.As illustrated by the schematic representation 1900 in FIG. 19, theup-sampled DMT signal 1910 may be hard clipped at 1902. Then a 2N-pointFFT is performed at 1904. The high-order bins N to 2N−1 are set to zeroto effect the OOB filtering at 1906. This is followed by a 2N-point IFFT1908. The method performs well, but the additional 2N-point FFT/IFFT1908 operation may incur unacceptably high hardware costs and/orprocessing burdens.

Note that the frequency domain filtering operation 1900 in FIG. 19 isgiven by the point-wise multiplication with the brick wall function

$H_{k}^{(c)} = \left\{ \begin{matrix}1 & {{k = 0},1,\ldots \mspace{14mu},{N - 1}} \\0 & {{k = N},\ldots \mspace{14mu},{{2\; N} - 1}}\end{matrix} \right.$

This operation is equivalent to circular convolution with the sincfunction

h _(n) ^((c)) =F[H ^((c))] where F is the DFT operator

This inspires an essentially equivalent method that operates in the timedomain utilizing circular convolution with h_(n) ^((c)), eliminating theneed for the additional FFT/IFFT. The significant reduction of OOB clipnoise is shown in the comparison of the plots of FIG. 18. It will beappreciated that the clipping may be set to provide about 12 dB ofheadroom for the transmitted DMT signal. Module 730 performsinterpolation to DAC rate. The half-band output may be sent to a Farrowinterpolator, which increases the sampling rate from 2F_(s) to2F_(c)=2F_(s)×64/63.

Data Framing

Data framing affects the operation of both SLOC transmitter and SLOCreceiver. A superframe and an interleave-frame are defined herein. Asuperframe structure may be distinct from the interleave-frame that isused for upstream signals. A downstream superframe comprises asuperframe DMT sync symbol, a DMT system data (sysdata) symbol with nopilot subcarriers, and 708 DMT data symbols, each with a pair ofrotating pilot subcarriers. An upstream superframe comprises asuperframe DMT sync symbol, 2 DMT sysdata symbols each with a rotatingpilot subcarrier, and 352 DMT data symbols, each with a rotating pilotsubcarrier. The superframe can be viewed as containing 355 packages. Dueto the half-duplex alternating format of two downstream DMT symbols thenone upstream DMT symbol per package, the upstream and downstreamsuper-frames may coincide. This is illustrated in the timing diagrams2000 of FIG. 20.

A DMT superframe sync symbol (downstream and upstream) may comprise aspecific uncoded PN sequence in the form of BPSK symbols modulating thedata subcarriers. The four sync subcarriers are also present. Thissequence can be robustly detected by the receiver using means known inthe art. In one example, the start of a superframe is used to:

-   -   Synchronize the transmitter randomizer and the receiver        derandomizer. At the start of the super-frame the registers are        set to the value F180H for downstream, 3180H for upstream.    -   Synchronize the transmitter moving pilot pattern with the        expected pilot locations at the receiver digital equalizer. All        downstream pilot subcarriers are covered after exactly 354        downstream DMT data symbols. All upstream pilot subcarriers are        covered after exactly 2 upstream sysdata symbols and 175        upstream DMT data symbols. Thus there are two full rotations or        circulations per superframe.    -   Indicate to the receiver the location of the DMT sysdata        symbols.

As noted, certain embodiments employ a rotating pilot pattern. For eachdownstream DMT symbol, a different pair of BPSK modulated pilotsubcarriers may be assigned within the useful subcarrier range of510-1923. These subcarriers pairs may be assigned in a rotating manner.In one example, only even-numbered subcarriers are assigned except forsubcarrier 1923. All pilot subcarriers can be covered after 354 DMTsymbols (i.e., one super-frame). For example a process for pilotsubcarrier assignment may operate as follows:

-   -   f_(l)=510, f_(u)=1923;

at superframe sync:

-   -   initialize k=0;    -   at start of each DMT symbol:    -   N=0;    -   while N<2        -   BR_count=bit-reversed value of k;        -   if {(f_(l)≦BR_count≦f_(u)) && (BR_count is even)} or            -   (BR_count=1923)        -   then            -   BR_count is a pilot subcarrier for this DMT symbol;            -   N=N+1;        -   k=k+1;

A similar process may be adopted for the upstream DMT signal, although afew differences may be evident. One difference relates to the “whileloop” condition, which may be changed to N<1 so that only one pilotsubcarrier is assigned per DMT symbol. The bounds (f_(l)=63, f_(u)=239)may be different and both even and odd subcarriers may be assigned aspilots. Thus:

-   -   f_(l)=63, f_(u)=239;

at superframe sync:

-   -   initialize k=0;    -   at start of each DMT symbol:    -   N=0;    -   while N<1        -   BR_count=bit-reversed value of k;            -   if {(f_(l)≦BR_count≦f_(u))}            -   then                -   BR_count is a pilot subcarrier for this DMT symbol;                -   N=N+1;            -   k=k+1;

Upstream DMT sysdata symbols may structured as follows:

-   -   8 bytes of sysdata are carried per superframe    -   DMT sysdata symbol #1        -   4 bytes-16 uncoded 4-QAM symbols in 4 subbands        -   Data repeated 11 times over all 44 subbands    -   DMT sysdata symbol #2        -   4 bytes-16 uncoded 4-QAM symbols in 4 subbands        -   Data repeated 11 times over all 44 subbands

Downstream DMT sysdata symbols are structured as follows:

-   -   8 bytes of sysdata are carried per superframe    -   DMT sysdata symbol        -   8 bytes-32 uncoded 4-QAM symbols in 1 subband        -   Data repeated 44 times over all 44 subbands

The 8 bytes of data contain the following elements:

-   -   subband number (1-44)    -   constellation assigned to that subband (indicated by numbers 0-7        respectively representing null, 4-QAM, 8-PSK, 16-QAM, 32-QAM,        64-QAM, 128-QAM, 256-QAM and 512-QAM).    -   transmitted power level associated with that subband    -   P parameter—number of bytes per RS packet    -   Interleaver B parameter        -   5-36 indicated by numbers 0-31 for upstream,        -   same as P for downstream    -   For upstream only, interleave-frame DMT symbol count        (xmitter_W_count)

It will be appreciated that the subband number, constellationassignment, subband power level, and B and P parameters are transmittedfor use by the opposite side transmitter. The element xmitter_W_count istransmitted for use by the opposite side receiver.

Based on a current channel estimate, the upstream receiver may read thesysdata from the 4 subbands having the highest SNR for each of therespective four parts of the sysdata. These subbands need not beadjacent. For the downstream signal, the downstream receiver may readthe sysdata from one subband having the highest SNR.

Further robustness for the received sysdata can be achieved by using aconfidence counter (conf_ctr) algorithm, which may be implemented as aconf_ctr element in any combination of hardware and software for eachelement of the sysdata information. This is illustrated in the flowchart2100 of FIG. 21 which may apply to all sysdata elements except theupstream xmitter_W_count. In the downstream and upstream receivers theremay be 46 separate conf_ctr elements. These cover the constellationassignments and power levels for the 44 subbands and the B and Pparameters. In one example, values for flow chart parameters are max=16and thresh=6.

In the upstream receiver there may be an additional conf_ctr element forxmitter_W_count, as discussed elsewhere herein.

The upstream interleave-frame may be defined to include W DMT datasymbols, where W is determined by means described elsewhere herein. Theinterleave-frame start is not usually aligned to the superframe start.Included in the upstream DMT sysdata symbol subsequent to each DMTsuperframe sync symbol is an element containing the current transmitterinterleave-frame DMT data symbol count value (xmitter_W_count). This isa modulo W counter. Note that for the downstream signal, W is typicallyeffectively set to 1.

Unlike the other sysdata elements, xmitter_W_count is not just receivedand saved. The receiver has its own interleave frame symbol counter,rcvr_W_count, that may be incremented modulo W as each DMT data symbolis received. An algorithm, such as the algorithm illustrated in theflowchart 2200 of FIG. 22 enables synchronization of the transmitter andreceiver counters. This results in robust interleave-framesynchronization so that the upstream deinterleaver correctlydeinterleaves the received bytes. The significance of the W parameter isexplained elsewhere herein.

Data Mapping to DMT Data Symbols

There can be exactly 1408 data subcarriers per downstream DMT symbol,and 1408/8=176 data subcarriers per upstream DMT symbol and subcarriersmay be modulated with QAM symbols. The number of net data bits carriedby each data subcarrier depends on the constellation assigned to the QAMsymbols for that subcarrier (see Table 2).

TABLE 2 Constellation m bits per m − 1 (net) bits per (2^(m)) QAM-symbolQAM-symbol  8 PSK 3 2  16 QAM 4 3  32 QAM 5 4  64 QAM 6 5 128 QAM 7 6256 QAM 8 7 512 QAM 9 8

As a result of this bit-loading, the net number of transmitted data bitsper DMT symbol (and thus the net bit rate) can vary with the selectedconstellation-to-subband mappings. Regardless of the selectedbit-loading assignments, it may be desirable to have a consistentalignment between RS packet start points and DMT symbol boundaries. Thiscan be accomplished by having an exact integral number of RS packets perV DMT symbols, where V is a small integer.

In some embodiments, (i) The number of RS packets per DMT symbol is U/Vwhere either U is an integer and V=1, or U=1 and V is an integer,regardless of the bit-loading assignments. It may also be desirable tohave consistent alignment between the top most commutator switch pointin the interleaver-deinterleaver and the DMT symbol boundaries. This canbe achieved by aligning the top switch position to every Wth DMT symbolboundary, where W denotes a given number of DMT symbols comprising aninterleave-frame. In some embodiments, (ii) the number of bytes in W DMTsymbols is exactly an integer multiple B, regardless of the bit-loadingassignments. Embodiments (i) provide for simple synchronization of theRS decoder in the receiver. The DMT receiver early processing willrobustly establish the location of DMT symbol boundaries by anyappropriate means known in the art. The location of DMT symbolboundaries, together with knowledge of V and P, can be used by RSdecoder to denote the RS packet starting points. Embodiments (ii)provide for simple synchronization of the receiver deinterleaver to thetransmitter interleaver. Knowledge of the DMT symbol boundary plus theDMT data symbol count within the length W “interleave-frame” can be usedby the receiver to synchronize the deinterleaver commutator switchposition to match that of the transmitter interleaver. This count ismaintained in the receiver and synchronized to the corresponding countin the transmitter.

Corollaries to item (i) can be stated as follows: (1) the number ofbytes per V DMT symbols must be an integer, and (2) the number of RSpackets per V DMT symbols must be an integer. The requirement stated inthe corollary 1 can be achieved by constraining the number ofsubcarriers per assignable subband to be an integer multiple of 4. Thisassures that each subband contains an integer number of half-bytesregardless of constellation assignment. See Table 3. In Table 3 it isindicated that an upstream (US) DMT symbol comprises 44 subbands of1×4=4 subcarriers each (44×4=176 data subcarriers), and a downstream(DS) DMT symbol consists of 44 subbands of 8×4=32 subcarriers each(44×32=1408 data subcarriers).

TABLE 3 m m−1 (net) 4 subcarriers 32 subcarriers Con- bits per bits per1 subband (US) 1 subband (DS) stella- QAM- QAM- net net net net tion(2^(m)) symbol symbol bits bytes bits bytes 8 PSK 3 2 8 1 64 8 16 QAM 43 12 1.5 96 12 32 QAM 5 4 16 2 128 16 64 QAM 6 5 20 2.5 160 20 128 QAM 76 24 3 192 24 256 QAM 8 7 28 3.5 224 28 512 QAM 9 8 32 4 256 32

For the upstream signal, it can be seen that because some subbands maycarry an integer plus one-half bytes, two DMT symbols are neededguarantee an integer number of bytes. For the downstream signal, asingle DMT symbol will always carry an integer number of bytes.

To meet the requirement of corollary 2, P may be allowed to vary withthe bit-loading (the number of parity bytes may be kept constant withonly the number of data bytes varying). For the upstream signal, V=2,U=1; U/V=½ RS packets per DMT symbol, 2 DMT symbols carry exactly 1 RSpacket. For the downstream signal, V=1, U=4; U/V=4 RS packets per DMTsymbol.

Item (ii) provides for the simple synchronization of the interleaver anddeinterleaver. For the downstream signal this is easily achieved asfollows:

-   -   B parameter (B_(DS)) set equal to P    -   M_(DS)=1    -   W_(DS)=1

Thus the number of bytes per W=1 DMT symbol is exactly 4B_(DS).

As for the downstream signal, it may be desired that the number of bytesin W DMT symbols be an integer multiple B_(US). Achieving this for theupstream signal is somewhat more complicated and is explained asfollows:

-   -   B_(US)=ceil(B_(DS)/6),M_(US)=6    -   Since an upstream DMT symbol may include a half-byte, set        W_(US)=even integer×B_(US)

For SLOC upstream, Wus=2×B_(US). This defines the length of an upstreaminterleave-frame. The upstream interleave-frame boundaries aredetermined at the upstream receiver. In the following two examples, itis assumed that the channel response measured by the respective upstreamand downstream receivers is virtually the same. In that case thebit-loading assignments will be spectrally identical and P_(US)=P_(DS).

However, that need not be the case. In the first example, the upstreamand downstream constellation to subcarrier assignments are given inTable 4 (upstream constellation mappings for 100.8 Mbps downstreamoperation over 2000 feet of RG-59) and Table 5 (downstream constellationmappings for 100.8 Mbps downstream operation over 2000 feet of RG-59),respectively and illustrated in FIG. 3.

TABLE 4 US bytes per net bit-rate Constellation # subbands # subcarriersDMT sym Mbps  8 PSK 7 28 7 0.43  16 QAM 9 36 13.5 0.83  32 QAM 6 24 120.74  64 QAM 6 24 15 0.92 128 QAM 6 24 18 1.10 256 QAM 6 24 21 1.29 512QAM 4 16 16 0.98 totals 44 176 102.5 6.29

TABLE 5 DS bytes per net bit-rate Constellation # subbands # subcarriersDMT sym Mbps  8 PSK 7 28 × 8 = 224 7 × 8 = 56 6.88  16 QAM 9 36 × 8 =288 13.5 × 8 = 108 13.28  32 QAM 6 24 × 8 = 192 12 × 8 = 96 11.80  64QAM 6 24 × 8 = 192 15 × 8 = 120 14.75 128 QAM 6 24 × 8 = 192 18 × 8 =144 17.70 256 QAM 6 24 × 8 = 192 21 × 8 = 168 20.65 512 QAM 4 16 × 8 =128 16 × 8 = 128 15.74 totals 44 1408 102.5 × 8 = 820 100.80

This bit-loading is provided to enable the 100 Mbps downstream bit rateto be achieved, given the cable frequency response of FIG. 1. Other bitrate targets combined with different cable types and/or lengths may usedifferent constellation-to-subcarrier mappings. In this example, eachupstream DMT symbol carries 102.5 bytes. Two DMT symbols carry 205 byteswhich is P. For the downstream signal, each DMT symbol carries 820bytes. P is 820/4=205 bytes (same for upstream and downstream). For theinterleaver-deinterleaver we have

-   -   P_(DS)=205; P_(US)=205    -   B_(DS)=P_(DS); B_(US)=ceil(P_(US)/6)=ceil(34.167)=35    -   W_(US)=2×35=70 DMT data symbols per interleave-frame    -   number of bytes in 70 upstream DMT data symbols is 70×102.5=7175    -   Requirement is that 7175 is an integer multiple of 35:        7175/35=205

The impulse duration tolerance goal is one package as illustrated inFIG. 23. The respective upstream and downstream impulse durationtolerance is:

-   -   B_(US)×t=35×12 bytes=2.049 RS packets >1 package.    -   B_(DS)×t=205×12 bytes=12 RS packets >1 package.

The second example addresses the case of 3000 feet of RG-59 cable. Thefrequency response for 3000 feet of RG-59 is shown in FIG. 24.Approximately 30 dB of tilt is observed across the SLOC signaling band.The upstream and downstream constellation to subcarrier assignments aregiven in Table 6 (upstream constellation mappings for 52.29 Mbpsdownstream operation over 3000 feet of RG-59) and Table 7 (downstreamconstellation mappings for 52.29 Mbps downstream operation over 3000feet of RG-59), respectively and illustrated in FIG. 25.

TABLE 6 US bytes per net bit-rate Constellation # subbands # subcarriersDMT sym Mbps  8 PSK 3 12 3 0.17  16 QAM 5 20 7.5 0.41  32 QAM 5 20 100.55  64 QAM 5 20 12.5 0.69 128 QAM 4 16 12 0.66 256 QAM 4 16 14 0.77512 QAM 0  0 0 0.00 totals 26 (of 44) 104 (of 176) 59 3.26

TABLE 7 DS bytes per net bit-rate Constellation # subbands # subcarriersDMT sym Mbps  8 PSK 3 12 × 8 = 96 3 × 8 = 24 2.66  16 QAM 5 20 × 8 = 1607.5 × 8 = 60s 6.65  32 QAM 5 20 × 8 = 160 10 × 8 = 80 8.86  64 QAM 5 20× 8 = 160 12.5 × 8 = 100 11.08 128 QAM 4 16 × 8 = 128 12 × 8 = 96 10.63256 QAM 4 16 × 8 = 128 14 × 8 = 112 12.41 512 QAM 0 0 × 8 = 0 0 × 8 = 00.00 totals 26 (of 44) 823 (of 1408) 59 × 8 = 472 52.28

This second example illustrates operation over a 3000 foot cable wherethe receiver SNRs for the higher frequency subbands would be too low forreliable reception due to the increased high-frequency attenuation ofthe longer cable. The higher frequency subbands are therefore not used.This reduces the net bit rate but allows for reliable reception of dataat the lower rate. Note that due to the nulling of the higher order datasubcarriers, we can transmit the active subcarriers with more power persubcarrier. Each upstream DMT symbol carries 59 bytes. Two DMT symbolscarry 118 bytes which is P. For the downstream signal, each DMT symbolcarries 472 bytes. P=472/4=118 bytes (same for upstream and downstream).For the interleaver-deinterleaver we have:

-   -   P_(DS)=118; P_(US)=118    -   B_(DS)=P_(DS); B_(US)=ceil(P_(US)/6)=ceil(19.67)=20    -   W_(US)=2×20=40 DMT data symbols per interleave-frame    -   number of bytes in 40 upstream DMT data symbols is 40×59=2360    -   Requirement is that 2360 is an integer multiple of 20:        2360/20=118

The impulse duration tolerance goal is one package as illustrated inFIG. 23. The respective upstream and downstream impulse durationtolerance is

-   -   B_(US)×t=20×12 bytes=2.034 RS packets >1 package    -   B_(DS)×t=205×12 bytes=12 RS packets >1 package

Table 8 shows the range of values of P that result from the bit-loadingcalculations over a wide range of cable lengths. Also shown are theB_(DS), B_(US), W_(US) and impulse tolerance in RS packets for each P.

TABLE 8 Downstream, M = 1 Upstream, M = 6 max impulse max impulseB_(DS)t/P B_(US) = B_(US)t/P W_(US) = P_(DS), P_(US) B_(DS) = P_(DS)packets ceil(P_(US)/6) packets 2 × B_(US) 211-216 211-216 12 362.05-2.00 72 205-210 205-210 12 35 2.05-2.00 70 199-204 199-204 12 342.05-2.00 68 193-198 193-198 12 33 2.05-2.00 66 187-192 187-192 12 322.05-2.00 64 181-186 181-186 12 31 2.06-2.00 62 175-180 175-180 12 302.06-2.00 60 169-174 169-174 12 29 2.06-2.00 58 163-168 163-168 12 282.06-2.00 56 157-162 157-162 12 27 2.06-2.00 54 151-156 151-156 12 262.07-2.00 52 145-150 145-150 12 25 2.07-2.00 50 139-144 139-144 12 242.07-2.00 48 133-138 133-138 12 23 2.08-2.00 46 127-132 127-132 12 222.08-2.00 44 121-126 121-126 12 21 2.08-2.00 42 115-120 115-120 12 202.09-2.00 40 109-114 109-114 12 19 2.09-2.00 38 103-108 103-108 12 182.10-2.00 36  97-102  97-102 12 17 2.10-2.00 34 91-96 91-96 12 162.11-2.00 32 85-90 85-90 12 15 2.12-2.00 30 79-84 79-84 12 14 2.13-2.0028 73-78 73-78 12 13 2.14-2.00 26 67-72 67-72 12 12 2.15-2.00 24 61-6661-66 12 11 2.16-2.00 22 55-60 55-60 12 10 2.18-2.00 20 49-54 49-54 12 92.20-2.00 18 43-48 43-48 12 8 2.23-2.00 16 37-42 37-42 12 7 2.27-2.00 1431-36 31-36 12 6 2.32-2.00 12 26-30 26-30 12 5 2.30-2.00 10

Start-Up

System start-up begins with the repeated transmission of two PNsequences (tr0 and tr1) in the form of BPSK symbols modulating the datasubcarriers. For downstream, the sequences are based on a length-2047binary PN sequence shortened to length-1414. For upstream, the sequencesare based on a length-255 sequence shortened to length-177. Thesesequences assist the receiver in achieving sample synchronization andDMT symbol synchronization. Also, the receiver may use the training datato calculate an accurate channel estimate and SNR per subband estimate.The channel estimate may in turn be used to determine the initialfrequency domain equalizer tap weights using methods known in the arts.

Channel and SNR estimates may then be fed by the receiver to abit-loading (BL) algorithm that calculates for the opposite sidetransmitter the desired:

-   -   constellation to subband assignments    -   subband transmit power levels    -   number of bytes per DMT data symbols, and the number of bytes        per RS packet (P)    -   B_(US) and W_(US) for upstream, B_(DS)=P, W=1 for downstream

As previously explained, the elements associated with the latter twobullet points are functions of the first bullet point. This sysdata isperiodically sent to the other side in DMT sysdata symbols for use bythe other side transmitter.

Each respective receiver may synchronize to the DMT superframe syncsymbols which may enable it to find the DMT sysdata symbols. The sysdatais reliably read from these symbols as described previously. When theconf_ctr elements reach threshold, the sysdata is used for thetransmitter:

-   -   constellation to subband assignments    -   subband transmit power levels    -   P, B, and W

Channel conditions may change slowly over time. The receivercontinuously updates the channel state and SNR per subband estimates.This may result in altered bit-loading parameters. Thus, the sysdata maychange and can be sent to the respective transmitter via the periodicDMT sysdata symbols. According to another aspect of the invention,system startup relates to the half-duplex TDMA synchronization betweenthe two sides. Both sides transmit according to predetermined protocolto avoid collision. This is managed on each side by using the sampleclock as a global clock to drive a counter that indicates to eachrespective side when to transmit its part of the package.

Framing Control Operation

With reference again to FIG. 7, a controller-pilot pattern generator(CPPG) module 718 accepts as input two sets of sysdata from the localreceiver. Input sysdata X 734 is received by the local receiver from theopposite side. Input sysdata X 734 can be used by the transmitter tocreate the framing for the signal it is to transmit. Input sysdata Y 736may be calculated by the local receiver and transmitted to the oppositeside in DMT sysdata symbols.

During start up, the CPPG 718, using the QAM symbol type input selector738 provided to the QAM symbol mux 716, selects the special input. TheCPPG 718 generates BPSK symbols for the PN sequence for tr0 for thefirst type of DMT training symbol, followed by BPSK symbols for the PNsequence tr1 for the second type of DMT training symbol. This isrepeated for a sufficient period of type to allow the receiver on theopposite side to achieve sample clock sync, DMT symbol sync, andequalization.

After start up, the CPPG 718 generates a superframe DMT sync symbolcoincident with every 710^(th) DMT symbol for the downstreamtransmitter, and every 355^(th) DMT symbol for the upstream transmitter.At that time BPSK symbols are output for the superframe DMT sync symbolcomprising a PN sequence and QPSK symbols for the current W_count(W_count to be used by opposite side receiver for interleave-framesync). This is fed to the QAM symbol mux 714 at the special input. Thisis followed by one DMT sysdata symbol with no pilot subcarriers(downstream), or two DMT sysdata symbols each with a single pilotsubcarrier (upstream), also fed to the QAM symbol mux at the specialinput. This sysdata will be used by the opposite side transmitter. Thesuperframe sync signal 740 may also be used to initialize the randomizer708.

The remainder of the DMT symbols of the superframe are DMT data symbols,which may comprise subcarriers modulated by either QAM data symbols fromthe mapper 714, null QAM symbols, rotating pilot or sync BPSK symbols.The respective inputs are selected by the CPPG 718 using signal 738based on the IFFT subcarrier to which the particular QAM/BPSK symbol isto be assigned. For the QAM data symbol coding, using signal 740 theCPPG 718 indicates to the PTCM coder 712 and mapper 714 the code rateand constellation to be used for the next QAM data symbol. This is basedon knowledge of the BL map from sysdata X 734.

As noted herein, subcarriers carrying pilots may be selected accordingto one or more algorithms. For the upstream signal, subcarriers 63-239may comprise 176 data carrying subcarriers and 1 pilot subcarrier with asubcarrier index that changes for each DMT data symbol. For a given DMTdata symbol, if the pilot is assigned to subcarrier x, the index for allthe data subcarriers at index x and above are increased by one so as to“make room” for the pilot subcarriers. For the downstream signal,subcarriers 519-1923 comprise 4 sync subcarriers at fixed indexes, 1408data subcarriers and 2 pilot subcarriers whose index changes for eachDMT data symbol. As with the upstream signal, the data subcarriers haveto “make room” for the pilot subcarriers. For a given DMT data symbol,it may be assumed that the pilots are assigned to index x and y, withx<y. The index for all data subcarriers at index x and above but belowindex y are incremented by 1. The index for all data subcarriers atindex y and above are incremented by 2.

The CPPG 718 feeds the RS coder 704 the RS packet size P based onsysdata X 734. It also sends the RS coder 704 a sync signal coincidentwith every Vth DMT data symbol (V=1 for downstream, V=2 for upstream.For the V=2 case, the count is synchronized to the superframe).

The CPPG 718 feeds the interleaver 706 the B parameter form sysdata X734 and a sync symbol coincident with every Wth DMT data symbol. Themodulo W count at every superframe DMT symbol time is loaded onto thatsuperframe DMT sync symbol as previously described.

With reference now to FIG. 26, at a receiver 2600, the BL map iscalculated in module 2616 based on the channel and SNR estimates fromthe digital frequency domain equalizer module 2614. The P, B, and Wparameters may be calculated based on the BL map as previouslydescribed. This is termed sysdata Y and is a function of the channelconditions seen by this receiver. The sysdata Y 736 BL map incombination with superframe sync and DMT symbol sync may be used by thesoft demapper/Viterbi decoder module 2620 to know the constellation andcode rate for current input the QAM symbol. The W and P parameters arefed respectively to the deinterleaver 2624 and RS decoders 2626.

The control module 2618 is fed by superframe sync and DMT symbol sync.From this information it is known whether the QAM symbols at thedemultiplexer (demux) input are DMT data symbols or DMT sysdata symbols.This information is used to control the demux 2630 so as direct the QAMsymbols to either the soft demapper/Viterbi decoder 2620 or the readsysdata module 2628.

The read sysdata module extracts sysdata X and executes the confidencecounter algorithm of FIG. 21. This is fed to the local transmitter. Forthe W_count, the algorithm of FIG. 22 is executed and this value is fedback to the control module which uses it to update a modulo W DMT datasymbol counter used to generate the sync at Wth DMT data symbol signal.

Based on the sysdata Y input and the input sync signals, the controlmodule outputs a sync signal every Wth DMT symbol to synchronize thedeinterleaver operation. It also outputs a sync signal every Vth DMTsymbol to synchronize the RS decoder operation.

SLOG DMT Receiver

With continued reference to the DMT receiver of FIG. 26, an ADC runningat nominally 90 MHz provides input to a digital AGC and HPF module 2604.The AGC provides 12-13 dB of headroom while the HPF suppresses theremaining CVBS signal after the analog BPF in the receiver AFE. Sampleclock frequency and phase recovery as well as DMT symbol timing recoveryare briefly described here.

The DMT system may be very sensitive to sample clock frequency and phaseerror. Since the DMT transmitter and receiver each have their own localoscillators, frequency error may initially exist. At the camera side,the DMT transmitter and DMT receiver use the same sample clock. If theDVR side receiver can synchronize to the camera side transmitter sampleclock, and then use that clock for its transmitter, all four sampleclocks will be synchronized.

A high-level view 2700 of the sample clock scheme and/or generator isshown in FIG. 27. The RC blocks are sample rate converters that utilizeFarrow architectures to achieve sample rate conversion. As can be seen,only the DVR side receiver need utilize a synchronization loop.

The rate converter control loop 2720 for the DVR side DMT receiver isshown in FIG. 27. The sample clock frequency and phase error areestimated using frequency domain data. The error estimates are used bythe rate convertor to adjust the interpolation timing using well knownmethods. Frequency synchronization is achieved during system start-upwith the aid of DMT training symbols during start-up. The phase istracked during normal data transmission with the aid of the four pilotsubcarriers.

Also during system start-up, at the receiver the boundaries of thetransmitted DMT symbols must be found. This is typically needed tosynchronize the FFT starting point so as to preventinter-symbol-interference. In the receiver a time domaincross-correlation operation using the a priori known transmitted PNsequence finds the peaks from which the DMT symbol boundaries areinferred.

The receiver may employ Fast Fourier Transforms (FFTs). Let r[m] be themth N-length vector of DMT signal samples at the input to the N-pointFFT module of FIG. 31. That vector is given by:

${r\lbrack m\rbrack} = {\left\lbrack {{r_{\frac{- N}{2}}\lbrack m\rbrack}{r_{\frac{- N}{2} + 1}\lbrack m\rbrack}\mspace{14mu} \ldots \mspace{14mu} {\ldots.\mspace{14mu} {r_{\frac{N}{2} - 1}\lbrack m\rbrack}}} \right\rbrack.}$

Then the FFT efficiently computes the vector

${Á\lbrack m\rbrack} = \left\lbrack {{Á\lbrack m\rbrack}{Á_{\frac{- N}{2} + 1}\lbrack m\rbrack}\mspace{11mu} \ldots \mspace{14mu} {\ldots.\mspace{14mu} {Á_{\frac{N}{2} - 1}\lbrack m\rbrack}}} \right\rbrack$where${{Á_{k}\lbrack m\rbrack} = {\frac{1}{\sqrt{N}}{\sum\limits_{n = {{- N}/2}}^{{N/2} - 1}\; {{r_{n}\lbrack m\rbrack}^{\frac{{- j}\; 2\; \pi \; {kn}}{N}}}}}},{k = \frac{- N}{2}},{\frac{- N}{2} + 1},\ldots \mspace{14mu},{\frac{N}{2} - 1.}$

In order to compute the FFT, the DMT receiver employs a pipelinedarchitecture composed of log₂N stages of “butterfly boxes”, eachincluding a complex multiplier. (The transmitter IFFT architecture issimilar.) Memory requirements are on the order of N to 2N and varysomewhat with the particular architecture chosen. For the downstreamreceiver, the 4096-point decimation in frequency (DIF) FFT modulecomprises six radix-4 stages. For the upstream receiver, the 512-pointDIF FFT module comprises one radix-2 stage followed by four radix-4stages. The DIF FFT input is in normal subcarrier order. The output isin bit-reversed subcarrier order.

Channel Estimation and Equalization

After AGC and synchronization have been achieved, the pilot subcarriersprovide the channel estimator with the information it needs to calculatean estimate of the cable frequency response. During training, all theuseful subcarriers can be considered to be pilots. After training, thepilot subcarriers rotate or circulate as previously explained. Theinstantaneous channel estimate at DMT symbol time m is:

Ĥ[m]=[Ĥ _(f) _(l) [m]Ĥ _(f) _(l+D) [m] . . . Ĥ _(f) _(l+M D) Ĥ _(f) _(u)[m]]

where f_(l) . . . f_(u) is the span of useful subcarriers (63-239upstream, 510-1923 downstream). For the downstream signal, M=706 andD=2. For the upstream signal, M=175 and D=1. After training, at each DMTsymbol time m, only p elements of the vector are updated; the restretain their values computed during previous DMT symbols. Fordownstream, p=2. For upstream, p=1. Thus each element of the complexvector H is determined by the magnitude and phase of the received BPSKsymbol when the corresponding subcarrier is assigned to be a pilot. Eachelement (frequency bin) of the vector is given by:

${{{\hat{H}}_{i}\left\lbrack q_{i} \right\rbrack} = \frac{Z_{i}\left\lbrack q_{i} \right\rbrack}{P_{i}\left\lbrack q_{i} \right\rbrack}},{i = f_{l}},f_{l + D},{\ldots \mspace{14mu} f_{l + {MD}}},f_{u}$

where Z_(i) is the received sample, P_(i) is the a priori known BPSKpilot symbol, and i is a pilot subcarrier number during DMT symbol timeq_(i).

The rotating pilots allow for the tracking of slow changes to thechannel frequency response. The frequency bins are updated over time by:

{tilde over (H)} _(i) [q _(i)]=(1−β)Ĥ _(i) [q _(i) ]+β{tilde over (H)}_(i) [q _(i) −j]

where β is a forgetting factor and j is 177 for upstream and 354 fordownstream, i.e. the time relative to q_(i) when the pilot forsubcarrier i was last received.

Due to the smaller IFFT for the upstream signal, its frequency bins arerelatively wide. For that reason a pilot subcarrier is sent on arotating basis for all the subcarriers from f_(l) . . . f_(u) so that nointerpolation is needed to determine any of the subcarrier channelestimates. For the downstream signal, with D=2, “in between” subcarrierchannel estimates are computed using cubic interpolation. The fullinterpolated channel estimate at time s, {acute over (H)}_(n) [s], isgiven by

$\mspace{20mu} {{{{For}\mspace{14mu} n} \in \left\{ {f_{l},f_{l + D},\ldots \mspace{14mu},f_{L + {MD}},f_{u}} \right\}},{{{\overset{\prime}{H}}_{n}\lbrack s\rbrack} = {{\overset{\sim}{H}}_{n}\left\lbrack q_{n} \right\rbrack}}}$$\mspace{20mu} {{{{For}\mspace{14mu} n} \in \left\{ {513,515,\ldots \mspace{14mu},1919} \right\}},{{{\overset{\prime}{H}}_{n}\lbrack s\rbrack} = {{\frac{- 1}{16}{{\overset{\sim}{H}}_{n - 3}\left\lbrack q_{n - 3} \right\rbrack}} + {\frac{9}{16}{{\overset{\sim}{H}}_{n - 1}\left\lbrack q_{n - 1} \right\rbrack}} + {\frac{9}{16}{{\overset{\sim}{H}}_{n + 1}\left\lbrack q_{n + 1} \right\rbrack}} + {\frac{- 1}{16}{{\overset{\sim}{H}}_{n + 3}\left\lbrack q_{n + 3} \right\rbrack}}}}}$$\mspace{20mu} {{{{For}\mspace{14mu} n} = 511},{{{\overset{\prime}{H}}_{511}\lbrack s\rbrack} = {{\frac{5}{16}{{\overset{\sim}{H}}_{510}\left\lbrack q_{510} \right\rbrack}} + {\frac{15}{16}{{\overset{\sim}{H}}_{512}\left\lbrack q_{512} \right\rbrack}} + {\frac{- 5}{16}{{\overset{\sim}{H}}_{514}\left\lbrack q_{514} \right\rbrack}} + {\frac{1}{16}{{\overset{\sim}{H}}_{516}\left\lbrack q_{516} \right\rbrack}}}}}$$\mspace{20mu} {{{{For}\mspace{14mu} n} = 1921},{{{\overset{\prime}{H}}_{1921}\lbrack s\rbrack} = {{\frac{1}{16}{{\overset{\sim}{H}}_{1916}\left\lbrack q_{1916} \right\rbrack}} + {\frac{- 5}{16}{{\overset{\sim}{H}}_{1918}\left\lbrack q_{1918} \right\rbrack}} + {\frac{15}{16}{{\overset{\sim}{H}}_{1920}\left\lbrack q_{1920} \right\rbrack}} + {\frac{5}{16}{{{\overset{\sim}{H}}_{1922}\left\lbrack q_{1922} \right\rbrack}.}}}}}$

Each received sample of DMT symbol s at the equalizer input (FFT output)is given by

Á _(n) [s]=H _(n) [s]A _(n) [s]+V _(n) [s] where V _(n) [s] is noise.

Assuming that the width of each subcarrier bin is small enough so thatthe channel response is approximately flat across each bin, zero-forcingequalization can be achieved by simply inverting the channel responsebased on the channel estimate. For DMT symbol s, the QAM symboltransmitted on subcarrier n where nε{f_(l) . . . f_(u)}, is estimated bythe equalizer as:

${Â_{n}\lbrack s\rbrack} = {{Á_{n}\lbrack s\rbrack} \cdot \frac{1}{{\overset{\prime}{H}}_{n}\lbrack s\rbrack}}$

where {acute over (H)}_(n) [r] is the most recent estimate of thechannel response for subcarrier n. This frequency domain multiplicationis equivalent to time domain circular convolution. However, the DMTcyclic suffix extension makes this effectively a linear convolution ofthe equalizer taps with the un-extended DMT symbol sample vector.

The equalizer output feeds PTCM coded quantized QAM data symbol samplesplus noise to the soft de-mapper which calculates soft bit-metrics usingwell-known algorithms. For correct operation, the soft de-mapper mustknow the constellation for the current QAM symbol. For this it may relyon DMT superframe symbol sync, DMT symbol sync and the bit-loadingassignment map (BL map) calculated in the calculate sysdata module 2616.

The soft bit metrics are fed to the Viterbi decoder. The Viterbialgorithm is well-known and executes soft decoding producing m−1 decodedbits per received QAM symbol. For correct operation, the Viterbi decodermust know the constellation for the current QAM symbol. For this itrelies on DMT superframe symbol sync, DMT symbol sync and thebit-loading (BL) assignment map calculated in the calculate sysdatamodule 2616. If the bit error rate at the Viterbi decoder output is2×10⁻⁴ or less, then the RS decoder output will be quasi-error-free.

The derandomizer has the same structure as the transmitter randomizer ofFIG. 9. The derandomizer may be synchronized to the superframe syncsignal from the control module 2618 and initialized as described herein.

The deinterleaver 850 of FIG. 8 inputs derandomized data bytes andrestores the original data byte ordering. It is synchronized by a signalfrom the control module 2618 that occurs at every Wth DMT data symbol.

The RS decoder determines the packet starting points based on a syncsignal every Vth DMT data symbol and knowledge of the packet size P. Thecorrected bit stream output from the Viterbi decoder is packed intobytes and fed to the SLOC RS decoder. A high-level flow diagram of theRS decoder is shown in FIG. 28. A syndrome calculator views the inputpacket as a polynomial with the bytes as GF(256) coefficients. Itcalculates 2t=12 syndromes. If all are zero, the input packet is a validcodeword. Otherwise the packet is corrupted by errors. Next, the keyequation solver determines the error locator and error value polynomialsusing, for example, the Berlekamp-Massey algorithm. A search may findthe corrupted bytes by evaluating the roots of the error locatorpolynomial. The error values are determined and the corrupted bytes arecorrected.

FIG. 29 shows details of the camera side AFE 2900 and DVR side AFE 2920of FIGS. 5 and 6, respectively. For the DMT Rx path, the filteringstages comprise a BPF with a variable gain stage inserted after thefirst filter stage. This BPF suppresses the near CVBS signal on the lowside. The DMT Tx path also uses a BPF to prevent interference into theCVBS on the low side, and image suppression on the high side (see FIG.30). In the CVBS path the LPF reduces interference into the DMT signal.At the DVR side, the DMT filtering is similar to that of the cameraside. For the CVBS path, the LPF suppresses the strong near DMT Tx.

Some IP cameras may not have a CVBS output and SLOC can take advantageof some of the available low-frequency spectrum space. Given the absenceof the requirement to transmit CVBS, SLOC may also utilize increased DMTtransmit power. Accordingly, the useful cable length may be extended at100 Mbps.

Turning now to FIG. 31, certain embodiments of the invention employ aprocessing system 3100 deployed to perform certain of the functionsdescribed herein. Processing system 3100 may comprise a commerciallyavailable system that executes commercially available operating systemssuch as Microsoft Windows®, UNIX or a variant thereof, Linux, a realtime operating system and or a proprietary operating system. Thearchitecture of the processing system may be adapted, configured and/ordesigned for integration in the processing system, for embedding in oneor more of an image capture system, a modem, a video processingworkstation, a DVR, video display system, video camera and/or a routeror other communications device. In one example, processing system 3100comprises a bus 3102 and/or other mechanisms for communicating betweenprocessors, whether those processors are integral to the processingsystem 3100 (e.g. processor 3104 or located in different, perhapsphysically separated processing systems 3100. Device drivers 3103 mayprovide output signals used to control internal and external components

Processing system 3100 also typically comprises memory 3106 that mayinclude non-transitory storage media such as random access memory(“RAM”), static memory, cache, flash memory, and any other suitable typeof storage device that can be coupled to bus 3102. Memory 3106 can beused for storing instructions and data that can cause one or more ofprocessors 3104 and 3105 to perform a desired process. Main memory 3106may be used for storing transient and/or temporary data such asvariables and intermediate information generated and/or used duringexecution of the instructions by processor 3104. Processing system 3100also typically comprises non-volatile storage such as read only memory(“ROM”) 3108, flash memory, memory cards or the like; non-volatilestorage may be connected to the bus 3102, but may equally be connectedusing a high-speed universal serial bus (USB), Firewire or other suchbus that is coupled to bus 3102. Non-volatile storage can be used forstoring configuration, and other information, including instructionsexecuted by processors 3104 and/or 3105. Non-volatile storage may alsoinclude mass storage device 3110, such as a magnetic disk, optical disk,flash disk that may be directly or indirectly coupled to bus 3102 andused for storing instructions to be executed by processors 3104 and/or3105, as well as other information.

Processing system 3100 may provide an output for a display system 3112,such as an LCD flat panel display, including touch panel displays,electroluminescent display, plasma display, cathode ray tube or otherdisplay device that can be configured and adapted to receive and displayinformation to a user of processing system 3100. Device drivers 3103 caninclude a display driver, graphics adapter and/or other modules thatmaintain a digital representation of a display and convert the digitalrepresentation to a signal for driving a display system 3112. Displaysystem 3112 may also include logic and software to generate a displayfrom a signal provided by system 3100. In that regard, display 3112 maybe provided as a remote terminal, video monitor. For example, a modemmay process one or more signals representative of a video stream, wherethe one or more signals are transmitted over a coaxial cable. An inputdevice 3114 is generally provided locally or through a remote system andtypically provides for alphanumeric input as well as cursor control 3116input, such as a mouse, a trackball, etc. It will be appreciated thatinput and output can be provided to a wireless device such as a PDA, atablet computer or other system suitable equipped to display the imagesand provide user input.

According to one embodiment of the invention, portions of a SLOC modemmay be implemented by processing system 3100. Processor 3104 executesone or more sequences of instructions. For example, such instructionsmay be stored in main memory 3106, having been received from acomputer-readable medium such as storage device 3110. Execution of thesequences of instructions contained in main memory 3106 causes processor3104 to perform, or cause to be performed, process steps according tocertain aspects of the invention. In certain embodiments, functionalitymay be provided by embedded processing systems that perform specificfunctions wherein the embedded systems employ a customized combinationof hardware modules 3105 and software to perform a set of predefinedtasks. For example, customized hardware modules 3105 may perform certainsignal processing functions that would be difficult to implement insoftware executed on a processor 3104. Processor 3104 may comprise oneor more digital signal processors that perform certain operations onreceived signals. Accordingly, embodiments of the invention are notlimited to any specific combination of hardware circuitry and software.

The term “computer-readable medium” is used to define any medium thatcan store and provide instructions and other data to processor 3104and/or 3105, particularly where the instructions are to be executed byprocessor 3104 and/or 3105 and/or other peripheral of the processingsystem. Such medium can include non-volatile storage, volatile storageand transmission media. Non-volatile storage may be embodied on mediasuch as optical or magnetic disks, including DVD, CD-ROM and BluRay.Storage may be provided locally and in physical proximity to processors3104 and 3105 or remotely, typically by use of network connection.Non-volatile storage may be removable from processing system 3104, as inthe example of BluRay, DVD or CD storage or memory cards or sticks thatcan be easily connected or disconnected from a computer using a standardinterface, including USB, etc. Thus, computer-readable media can includefloppy disks, flexible disks, hard disks, magnetic tape, any othermagnetic medium, CD-ROMs, DVDs, BluRay, any other optical medium, punchcards, paper tape, any other physical medium with patterns of holes,RAM, PROM, EPROM, FLASH/EEPROM, any other memory chip or cartridge, orany other medium from which a computer can read.

Transmission media can be used to connect elements of the processingsystem and/or components of processing system 3100. Such media caninclude twisted pair wiring, coaxial cables, copper wire and fiberoptics. Transmission media can also include wireless media such asradio, acoustic and light waves. In particular radio frequency (RF),fiber optic and infrared (IR) data communications may be used.

Various forms of computer readable media may participate in providinginstructions and data for execution by processor 3104 and/or hardwaremodules 3105, which may include sequencers and custom configured logic.For example, the instructions may initially be retrieved from a magneticdisk of a remote computer and transmitted over a network or modem toprocessing system 3100. The instructions may optionally be stored in adifferent storage or a different part of storage prior to or duringexecution.

Processing system 3100 may include a communication interface 3118 thatprovides two-way data communication over a network 3120 that can includea local network 3122, a wide area network or some combination of thetwo. For example, an integrated services digital network (ISDN) may usedin combination with a local area network (LAN). In another example, aLAN may include a wireless link. Network link 3120 typically providesdata communication through one or more networks to other data devices.For example, network link 3120 may provide a connection through localnetwork 3122 to a host computer 3124 or to a wide area network such asthe Internet 3128. Local network 3122 and Internet 3128 may both useelectrical, electromagnetic or optical signals that carry digital datastreams.

Processing system 3100 can use one or more networks to send messages anddata, including program code and other information. In the example ofthe Internet, a server 3130 may transmit a requested code for anapplication program through Internet 3128 and may receive in response adownloaded application that provides for the anatomical delineationdescribed in the examples above. The received code may be executed byprocessor 3104 and/or 3105.

FIG. 32 includes a flowchart 3200 of a method of communication accordingto certain aspects of the present invention. The method may be performedin a modem, and various elements of the modem may comprise a computerprocessor, a digital signal processor, one or more sequencers, signalprocessors, field programmable devices, application specific integratedcircuits and/or dedicated logic. Various functional modules and/or otherelements may perform one or more steps of the method. Certain functionalelements are illustrated and described in regard to FIGS. 5-9, 19 and26-29, and these modules may individually comprise combinations of themodem and software modules.

At step 3202, the modem receives DMT symbols comprising a plurality ofQAM symbols. Each QAM symbol may modulate a subcarrier of a subband in areceived signal. Each QAM symbol may be one of a constellation of QAMsymbols assigned to the subband. The QAM constellation may be one of aplurality of QAM constellations assignable to the subband.

At step 3204, the modem decodes data carried by the QAM symbols. Thedata carried by the QAM symbols may be decoded using a block errordetecting decoder, such as a Reed-Solomon decoder. The Reed Solomondecoder (or other block decoder) may be used to decode a first integernumber (V) of code words which may be referred to herein as code blocks,and/or Reed-Solomon packets. For example, Reed-Solomon packets may beencoded in a second integer number (U) of the DMT symbols. U and V maybe integers. The Reed-Solomon decoder may be configured to besynchronized at the common boundary regardless of which of the pluralityof QAM constellations is assigned to the subband. A common boundary mayoccur at the start of each DMT symbol or at the start of eachReed-Solomon packet. V may be an integer multiple of U. A Reed-Solomonpacket may commences at the start of each DMT symbol. The Reed-Solomonpacket may commence at the start of each pair of the DMT symbols. Insome embodiments U is an integer multiple of V, and each Reed-Solomonpacket may commence at the start of a DMT symbol.

At step 3206, the Reed-Solomon decoder is synchronized at the commonboundary.

In some embodiments, the modem is configured to deinterleave bytes ofthe data carried by the QAM symbols using a deinterleaver synchronizedto a third integer number (W) of DMT symbols. In one example, the W DMTsymbols correspond to a fourth number of bytes of an interleave frameassociated with the received signal. The QAM constellation may be one ofa plurality of QAM constellations assignable to the subband.Deinterleaving may typically be performed by a deinterleaver configuredto be synchronized regardless of which of the plurality of QAMconstellations is assigned to the subband.

In some embodiments, the received signal comprises a plurality ofsubbands. Each of the plurality of subbands may include two or moreadjacent subcarriers of the received signal. Different QAMconstellations may be assigned to at least two of the plurality ofsubbands. In one example, lower-order constellations of QAM symbols maybe assigned to subbands that include higher-frequency subcarriers andhigher-order constellations of QAM symbols may be assigned to subbandsthat include lower-frequency subcarriers. A common QAM constellation maybe assigned to each of the plurality of subbands. A common constellationof QAM symbols may be assigned to a group of adjacent subbands.Different constellations of QAM symbols may be assigned to differentgroups of adjacent subbands. Two or more groups of adjacent subbands maycomprise different numbers subbands based on signal-to-noise ratiosassociated with the two or more groups of adjacent subbands.

In some embodiments, the received signal is received from a coaxialcable and a lowest-frequency subcarrier included in the plurality ofsubbands may have a higher frequency than a baseband video signaltransmitted through the coaxial cable. The DMT symbols may betransmitted through the coaxial cable in a subband of a transmittedsignal.

In some embodiments, the modem estimates channel quality in the coaxcable based on two or more pilots carried on a corresponding number ofsubcarriers in the received signal. Channel quality may include a signalto noise ratio of the coax cable, and may be related to phase shift andsusceptibility to impulse noise, etc. The two or more pilots may rotateor otherwise circulate between subcarriers of the received signal. Thetransmitted signal and the received signal may be transmitted inadjacent time intervals.

In some embodiments, two downlink DMT symbols may be received in each ofa plurality of successive downlink time intervals. Transmitting the DMTsymbols through the coaxial cable may include transmitting one uplinkDMT symbol in an uplink time interval that is defined after each of theplurality of successive downlink time intervals. A synchronization DMTsymbol may be received. Certain operational aspects of the modem may beconfigured using system configuration information received in each of aplurality of frames. Each frame may include a plurality of packages.Each package may include two downlink DMT symbols and one uplink DMTsymbol.

FIG. 32 includes a flowchart 3220 of a method of communication accordingto certain aspects of the present invention. The method may be performedin a modem, and various elements of the modem may comprise a computerprocessor, a digital signal processor, one or more sequencers, signalprocessors, field programmable devices, application specific integratedcircuits and/or dedicated logic. Various functional modules and/or otherelements may perform one or more steps of the method. Certain functionalelements are illustrated and described in regard to FIGS. 5-9, 19 and26-29, and these modules may individually comprise combinations of themodem and software modules.

At step 3222, the modem encodes data in an integer number (V) ofReed-Solomon packets.

At step 3224, the modem may select one or more subcarriers to carrypilot signals. The selection of pilot signals may be governed by themethods and algorithms disclosed herein. The modem may transmit two ormore pilots on a corresponding number of subcarriers in the downlinksignal, whereby the two or more pilots rotate or otherwise circulatebetween subcarriers of the downlink signal. In one example, the two ormore pilots rote between the subcarriers in a pattern that repeats foreach rotation. In another example, the two or more pilots may becontinuously or cyclically rotated according to a pattern that maycontinuously change.

At step 3226, the modem transmits the V Reed-Solomon packets in aninteger number (U) of

DMT symbols. Each DMT symbol may comprise a plurality of QAM symbolsthat modulate subcarriers of a subband in a downlink signal. Each QAMsymbol may be one of a constellation of QAM symbols assigned to thesubband. The transmission of each of the V Reed-Solomon packets may beinitiated coincident with a beginning of a DMT symbol.

In some embodiments, the modem interleaves bytes of the data using aninterleaver synchronized to a third integer number (W) of DMT symbols. Wmay be an integer.

In some embodiments, the modem selects the QAM constellation from aplurality of QAM constellations assignable to the subband. The modem mayselect the QAM constellation by assigning at least one of the pluralityof QAM constellations to one or more subbands based on signal-to-noiseratios associated with the one or more subbands, or based on some otherindicator of channel quality. The modem may transmit the downlink signalon a coaxial cable. A lowest-frequency subcarrier included in thedownlink signal may have a higher frequency than a baseband video signalcommunicated through the coaxial cable.

In some embodiments, the modem transmits two DMT symbols in each of aplurality of successive downlink time intervals, and receives one DMTsymbol from the coaxial cable in an uplink time interval that occursafter each of the plurality of successive downlink time intervals. Thus,the modem may utilize at least a portion of the available bandwidth in acommunication channel for half-duplex communications.

ADDITIONAL DESCRIPTIONS OF CERTAIN ASPECTS OF THE INVENTION

Certain embodiments of the invention provide systems and methods forcommunication involving video feeds. A method according to certainaspects of the invention comprises receiving a plurality of downlinksymbols modulated on a subband of subcarriers in a downlink signal. Afirst constellation of QAM symbols assigned to the subband may bedifferent from at least one other constellation of QAM symbols assignedto other subbands in the downlink signal. The method may comprisedecoding the plurality of downlink symbols using a block errorcorrection decoder. The block error correction decoder may besynchronized based on an identification of the first constellation ofQAM symbols and information identifying boundaries between the pluralityof downlink symbols.

In some embodiments, the subband is one of a plurality of subbands inthe downlink signal. Each of the plurality of subbands may comprise twoor more adjacent subcarriers of the downlink signal. Lower-orderconstellations of QAM symbols may be assigned to subbands that includehigher-frequency subcarriers. Higher-order constellations of QAM symbolsmay be assigned to subbands that include lower-frequency subcarriers.Each constellation of QAM symbols may be assigned to a group of adjacentsubbands. Two or more groups of subbands comprise different numbers ofgroups of subbands. The downlink signal may be received from a coaxialcable. A lowest-frequency subcarrier included in the plurality ofsubbands may have a higher frequency than a baseband video signaltransmitted through the coaxial cable. Constellations of QAM symbols maybe assigned to groups of one or more adjacent subbands based on asignal-to-noise ratio associated with the adjacent subbands of eachgroup of adjacent subbands.

In some embodiments, the downlink signal is received from a coaxialcable. The method may comprise transmitting a plurality of uplinksymbols through the coaxial cable in an uplink signal. An integer numberof bytes may be encoded in each of the plurality of uplink symbols. Aninteger number of bytes may be encoded in each of the plurality ofdownlink symbols. Each of the plurality of uplink symbols may encode adifferent number of bytes than the number of bytes encoded by each ofthe plurality of downlink symbols. The subband in the downlink signalmay be one of group of adjacent subbands. The group of adjacent subbandsmay comprise a first number of subcarriers. At least one of a pluralityof other groups of subbands in the uplink signal comprises a secondnumber of subcarriers. The first and second numbers may be different.

In some embodiments, the subband in the downlink signal comprises 32subcarriers and at least one subband in the uplink signal comprises 4subcarriers.

In some embodiments, the subband in the downlink signal carries two ormore downlink pilots on a corresponding number of subcarriers, wherein areceiver estimates channel quality based on the pilots. Channel qualitymay include signal to noise ratio measured in pilots by the receiver.Subcarriers carrying the two or more pilots may be selected according toa rotation.

In some embodiments, transmitting a plurality of uplink symbols throughthe coaxial cable includes selecting at least one subcarrier from asubband of the uplink signal to carry an uplink pilot. Transmitting aplurality of uplink symbols may include periodically selecting adifferent subcarrier from the subband of the uplink signal to serve asthe uplink pilot. An integer number of half bytes is encoded in the eachuplink symbol and an integer number of bytes is encoded in each downlinksymbol.

In some embodiments, transmitting the plurality of uplink symbolsincludes, for each of the plurality of uplink symbols, interleavingbytes of uplink data to obtain interleaved data, and encoding theinterleaved data in the each uplink symbol using a constellationassigned to the uplink subband. Interleaving bytes of uplink data mayinclude interleaving bytes of uplink data using a frame size determinedas a function of a constellation of QAM symbols assigned to the uplinksubband. A selected combination of power level and assignment ofconstellations of QAM symbols for each of the uplink and downlinksubbands may be selected in order to provide a spectral match betweenthe uplink and downlink subbands.

In some embodiments, the uplink signal and the downlink signal aretransmitted in adjacent time intervals. Two downlink symbols arereceived in each of successive downlink time intervals. Transmitting aplurality of uplink symbols through the coaxial cable may includetransmitting one uplink symbol after each of the successive downlinktime intervals. The method may comprise receiving a downlinksynchronization symbol and system configuration information in each of aplurality of frames. Each frame may include downlink intervals in whichdownlink symbols are received. Each of a plurality of frames maycomprise 355 packages, each package including two downlink symbols andone uplink symbol. In some embodiments, the method comprisestransmitting an uplink synchronization symbol in each of the pluralityof frames.

The foregoing descriptions of the invention are intended to beillustrative and not limiting. For example, those skilled in the artwill appreciate that the invention can be practiced with variouscombinations of the functionalities and capabilities described above,and can include fewer or additional components than described above.Certain additional aspects and features of the invention are further setforth below, and can be obtained using the functionalities andcomponents described in more detail above, as will be appreciated bythose skilled in the art after being taught by the present disclosure.

Although the present invention has been described with reference tospecific exemplary embodiments, it will be evident to one of ordinaryskill in the art that various modifications and changes may be made tothese embodiments without departing from the broader spirit and scope ofthe invention. Accordingly, the specification and drawings are to beregarded in an illustrative rather than a restrictive sense.

What is claimed is:
 1. A method of communication, comprising: encodingdata in a first integer number (V) of code blocks; and transmitting theV code blocks in a second integer number (U) of discrete multi-tonemodulation (DMT) symbols, each DMT symbol comprising a plurality ofquadrature amplitude modulation (QAM) symbols that modulate subcarriersof a subband in a downlink signal, wherein each QAM symbol is one of aconstellation of QAM symbols assigned to the subband, wherein a commonboundary occurs at a start of each DMT symbol or at a start of each codeblock.
 2. The method of claim 1, wherein transmitting the V code blocksincludes selecting the constellation of QAM symbols from a plurality ofQAM constellations assignable to the subband.
 3. The method of claim 2,wherein encoding data includes interleaving bytes of the data using aninterleaver synchronized to a third integer number (W) of DMT symbols.4. The method of claim 1, wherein selecting the constellation of QAMsymbols includes assigning at least one of a plurality of QAMconstellations to one or more subbands based on signal-to-noise ratiosassociated with the one or more subbands.
 5. The method of claim 1,wherein transmitting the V code blocks includes transmitting thedownlink signal on a coaxial cable, and wherein a lowest-frequencysubcarrier included in the downlink signal has a higher frequency than ahighest frequency assigned for transmitting a baseband video signalthrough the coaxial cable.
 6. The method of claim 5, whereintransmitting the V code blocks includes transmitting two or more pilotson a corresponding number of subcarriers in the downlink signal.
 7. Themethod of claim 6, wherein transmitting the two or more pilots includescirculating the two or more pilots through a plurality of subcarriersused by the downlink signal.
 8. The method of claim 1, whereintransmitting the transmitting the V code blocks includes: transmittingtwo DMT symbols in each of a plurality of successive downlink timeintervals; and receiving one DMT symbol from an uplink signal in anuplink time interval that occurs after each of the plurality ofsuccessive downlink time intervals.
 9. An apparatus comprising: atransmitter configured to transmit discrete multi-tone modulation (DMT)symbols, each transmitted DMT symbol comprising a plurality ofquadrature amplitude modulation (QAM) symbols, wherein each QAM symbolmodulates a subcarrier of a subband in a transmitted signal; and aprocessing system configured to assign a constellation of QAM symbols tothe subband in the transmitted signal based on a condition of a channelused by the transmitter, wherein each QAM symbol is one of theconstellation of QAM symbols.
 10. The apparatus of claim 9, wherein thetransmitter transmits a first integer number (V) of code blocks encodedin a second integer number (U) of the DMT symbols, wherein a commonboundary occurs at a start of each DMT symbol or at a start of each codeblock.
 11. The apparatus of claim 9, further comprising an interleaverconfigured to interleave bytes of data to be carried in the DMT symbols,wherein the interleaver is synchronized to a third integer number (W) ofDMT symbols.
 12. The apparatus of claim 9, wherein the condition of thechannel used by the transmitter comprises a signal-to-noise ratio. 13.The apparatus of claim 9, further comprising a receiver configured toreceive DMT symbols from a received signal, each received DMT symbolcomprising a plurality of QAM symbols of a second QAM constellation thatmodulate a subcarrier in a second subband; and a block error correctiondecoder synchronized to boundaries of the received DMT symbols, whereinthe transmitter and receiver operate at different data rates.
 14. Theapparatus of claim 13, wherein each of the plurality of QAM symbolscomprise a first integer number (V) of code blocks encoded in a secondinteger number (U) of the DMT symbols, wherein a common boundary occursat a start of each received DMT symbol or at a start of each code block,and wherein the block error correction decoder comprises a Reed-Solomondecoder synchronized to the common boundary.
 15. The apparatus of claim13, wherein the transmitted signal and the received signal arecommunicated through a coaxial cable, and further comprising a low passfilter configured to separate a baseband video signal from thetransmitted signal and the received signal.
 16. The apparatus of claim15, wherein the processing system configured to determine the conditionof the channel is based on two or more pilots in the received signal.17. The apparatus of claim 16, wherein the two or more pilots rotateamong subcarriers of the received signal.
 18. The apparatus of claim 13,further comprising a deinterleaver configured to deinterleave bytes ofthe data carried by the QAM symbols of the received DMT symbols, whereinthe deinterleaver is synchronized to a third integer number (W) of DMTsymbols, wherein the W DMT symbols correspond to a fourth number ofbytes of an interleave frame associated with the received signal.
 19. Acomputer-readable medium comprising code for: encoding data in a firstinteger number (V) of Reed-Solomon packets; and transmitting the VReed-Solomon packets in a second integer number (U) of discretemulti-tone modulation (DMT) symbols, each DMT symbol comprising aplurality of quadrature amplitude modulation (QAM) symbols that modulatesubcarriers of a subband in a downlink signal, wherein each QAM symbolis one of a constellation of QAM symbols assigned to the subband,wherein a common boundary occurs at a start of each DMT symbol or at astart of each code block.